The simulation of a microwave antenna made of nanofilm Ti3C2 (Mxene)

Hello! I am a student of the radio engineering faculty and now I am engaged in master's work. The purpose of my work is to find application for Ti3C2 in antenna technology. I found a lot of information from article http://advances.sciencemag.org/content/4/9/eaau0920

But I can't understand how to simulate the Ti3C2 dipole in CST Microwave Studio. I have been trying for five months to get the same results as in article (S11 values), but without the result. Apparently, I do not understand something in terms of building a model. Could you help me with it, please?
The fact is that I modeled the dipole in the CST Studio program with the parameters that I managed to get from the article. The remaining parameters were set based on pure common sense and logic…

Parameters:
The thickness of the nanofilm-1.4 µm
The length of the dipole – 0,0625 m
Frequency-2.4 GHz
The width of the dipole was set to 0.005 m

I considered a 100-µm-thick layer of lossy substrate as the PET sheet. The permittivity value is 3.1, the el. conductivity = 2,07e-3 S/m.

For the simulation the Ti3C2-layer was used a surface impedance model instead.

A discrete 50-ohm port is used as a source

Boundaries-open (add space).

The simulation was carried out in the time domain with an accuracy of -60 dB. As a result, the simulation S11 came out to -8,5dB, which does not correspond to the data obtained in the article (-36dB).

The version of CST Studio is 2017.

If you will not complicate, prompt, please, that I do not correctly. I had no previous experience with modeling nanomaterials. Maybe I need to improve the accuracy or reduce the step of the meshing. On the other hand, maybe I create a surface impedance model of a Ti3C2-layer incorrectly.

Screenshots and model are attached
Thanks in advance!

P.S. Sorry for my English!
 

Baluncore

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@Alekseykolesnik Welcome to PF.
Sorry for the delay. Your English is very good.

Unfortunately I do not have CST Microwave Studio.
I cannot open your pack.rar on this system, maybe use a screenshots.pdf file?

The dipole length you have is correct for free space.
Dielectric constant of substrate is 3.1 nearby, but is 1.00 far away = free space.
I wonder what is the effective dielectric constant and the velocity factor?

Can you do a frequency sweep to measure the resonant frequency of your model dipole?
 
Thank you for your answer!
Ok, I attached the screenshots in pdf.

I didn't fully understand what you said about the dielectric constant of the substrate. After all, the dipole is applied to the substrate, respectively, it is in close contact with the substrate. That is, the value 3.1 should be used, right?

I have a graph of S11 versus frequency, isn't that the frequency sweep?
 

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Last edited:

Baluncore

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I do NOT see any big problems with your model.
The far-field radiation pattern has deep nulls on the Y-axis. That is correct for a dipole.
The gain is a maximum of 2.23 dBi which is good, I would expect 2.15 dBi for a dipole.

Do you model the dielectric as polyester or as PET ?

I have a graph of S11 versus frequency, isn't that the frequency sweep?
That is correct. I can now see it in screenshots.pdf; Your result S11 shows the reflected energy is a minimum at a frequency of 2.35 GHz.

The simulation was carried out in the time domain with an accuracy of -60 dB. As a result, the simulation S11 came out to -8,5dB, which does not correspond to the data obtained in the article (-36dB).
The difference could be due to several things. Their signal generator may be better matched to their dipole, or their dipole may have higher resistive losses than yours.

To better match the dipole to the generator, plot the complex impedance of the dipole at the required frequency. Adjust the length of the dipole to reduce the reactance to zero. Adjust the width of the dipole to bring the resistance closer to 50 ohm. Repeat the process until S11 gets closer to their figure of -36dB.

That will also eliminate estimates of dielectric constant Er, and VF.
I didn't fully understand what you said about the dielectric constant of the substrate. After all, the dipole is applied to the substrate, respectively, it is in close contact with the substrate. That is, the value 3.1 should be used, right?
When a dipole is implemented as a resonator on a PCB with a ground plane, almost all the energy propagates in the dielectric between the ground plane and the dipole element. The velocity of that wave is less than c, by a velocity factor of vf = 1/√( Er ). It will not radiate efficiently with a ground plane.
However, you have NO ground plane, so the energy propagates around the element, mostly through free space with Er = 1.00, but also through the thin dielectric layer with Er = 3.1; Since the dielectric is thin, and only on one side, it plays only a small part. I would guess that in your model the combined Er of space and the PE is going to be less than Er = 1.1 Modelling capacitors or transmission lines with asymmetric dielectrics is difficult without FEM, but we can calculate an estimate for the effective Er.

We know the length of the half-wave dipole is 62.5 mm.
299.792458 / ( 2 * 0.0625 ) = 2398.34 MHz
We know centre frequency (from S11 minimum) is at 2.35 GHz.
Frequency ratio is 2.39834 / 2.35 = 1.02057
The square of 1.02057 is the effective Er = 1.04156

If you reduce the length of the dipole to 62.5 / 1.02057 = 61.24 mm the centre frequency will move back up to 2.4 GHz. That will improve the match and lower S11.

Notice that the presence of a dielectric near the dipole can have a significant effect on the velocity factor. Changes in design need to be monitored. An insulated wire dipole will be affected by the thickness and material of the insulation.
 
The gain is a maximum of 2.23 dBi which is good, I would expect 2.15 dBi for a dipole.
The authors of the invention told me in correspondence that the gain of a half-wave dipole can never be greater than 2.15 dBi. It is an axiom. That's what confuses me.

Do you model the dielectric as polyester or as PET ?
It seems to be the same in Russia.
It says here that PET belongs to the polyester family.

Thank you! Now I am optimizing the size of the dipole to obtain the required S11 (-36dB). I hope that I will succeed.

I thought that there are some features in the modeling of this type of Ti3C2 antenna as opposed to conventional metal antennas...
Maybe I need to improve the accuracy or reduce the step of the meshing, or I should use other boundary conditions.

This file contains sheet resistance for nanofilms of a certain thickness. Depending on the thickness of the nanofilm (and its resistance), the parameter S11 varies greatly. Maybe I need to set the ti3c2 material in the program as ohmic sheet with pure active resistance?
 

Attachments

I ran a parameterization of the length of the dipole in the CST programme and the value of S11 is less than -13dB failed to obtain.
 

Baluncore

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The authors of the invention told me in correspondence that the gain of a half-wave dipole can never be greater than 2.15 dBi. It is an axiom. That's what confuses me.
I agree. But if the dipole is longer than half wavelength it can have gain greater than a dipole. Also, I expect the error margin of FEM to be maybe 0.08 dB.
Try changing mesh to have more elements, does 2.23 dB change towards 2.15 dB.
Try changing dipole length to see if 2.23 dB changes towards 2.15 dB.

I ran a parameterization of the length of the dipole in the CST programme and the value of S11 is less than -13dB failed to obtain.
Impedance match must be correct resistive and correct reactive.
A dipole usually has Zo ≈ 70 ohms. But you drive it with 50 ohms.
Try changing the dipole width from x = 5 mm to x = 3 mm, or 8 mm, does impedance match improve and S11 become less ?
 
Impedance match must be correct resistive and correct reactive.
I have only the sheet resistance for Ti3C2-layer. The article does not specify the reactance, that's why I don't know how to set properties of Ti3C2-material in CST Studio rightly.

Changing the width of the dipole in the larger and smaller side leads to a shift in resonance at a frequency of 2.25 GHz, S11 does not change. The increase in impedance of the port to 70 Ω also leads to a shift of resonance frequency of 2.25 GHz and reduce S11 to -23dB.
 

Baluncore

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Changing the width of the dipole in the larger and smaller side leads to a shift in resonance at a frequency of 2.25 GHz, S11 does not change.
You must fix excitation frequency to 2.4GHz then match excitation impedance by changing the Mxene dipole dimension of larger and smaller sides.

I notice your substrate is same size as dipole. Maybe make the substrate bigger so you can keep it fixed during experiment. Here is a possible construction of model.

Make substrate sheet, 0.1 mm thick, below zero. Substrate; Polyester; Brick.
Xmin=-15e-3; Xmax=+15e-3; Ymin=-50e-3; Ymax=+50e-3; Zmin=-0.1e-3; Zmax=0;

Then print dipole on surface of substrate above zero. Dipole; Mxene; Brick.
Xmin = -5e-3; Xmax = +5e-3; Ymin = -31e-3; Ymax = +31e-3; Zmin = 0; Zmax = 1.4e-6;

Cut a 2 mm slot, through Mxene dipole for excitation feedpoint.
Do not cut the substrate; Vacuum; Brick.
Xmin = -15e-3; Xmax = +15e-3; Ymin = -1e-3; Ymax = +1e-3; Zmin = 0; Zmax = 1e-3;

Define excitation at the cut in Mxene dipole.
You must fix excitation at 2.4GHz.

Keep dimensions of substrate and vacuum cut fixed.
Adjust x and y dimension of Mxene to get (R+jX) ≈ 50 + j 0.

That will match dipole to 50 ohm excitation, S11 <= -36 dB, SWR = 1.0x?
Experiment with z dimension and properties of Mxene.
 
I did everything you said. As a result, I got the same S11 value as before (-13dB). I found the impedance graph (I attached it) but I don't know what exactly it shows. Maybe this graph shows the input impedance of the dipole?
Last Friday, I set the value of this schedule at a frequency of 2.4 GHz in the settings of the impedance of the port (~110 Ohm). So I was able to get the value S11=-37dB, which was necessary. Today I spent many hours, but I could not get the value S11=-36dB, but only S11=-26dB. I can't figure out if I made a mistake on Friday or if I'm doing something wrong now. I will try again to reach the value of S11=-36dB. Thank you for helping me! Maybe you know what this graph shows?
246627
 

Baluncore

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I believe the “discrete port impedance, (magnitude)” is the vector sum of the real and imaginary complex impedance components.
Z = ( R + j X ); Magnitude = √ ( R2 + X2 )

You need to plot port impedance as separate complex components, resistance R ohms, and reactance X ohms.
Then change the dipole ±x and ±y size to get Z = ( 50 + j 0 ) ohms at 2.4 GHz.

As substrate and vacuum gap are now much bigger than the dipole, you may ignore them while you;
Adjust dipole ±y to significantly change the reactance; imaginary = zero.
Adjust dipole ±x to significantly change the resistance; real = 50 ohm.
You can probably use component sweep to find optimum values of x and y.
 

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